# ISL6530EVAL2

Manufacturer Part NumberISL6530EVAL2
DescriptionEVALUATION BOARD 2 ISL6530
ManufacturerIntersil
ISL6530EVAL2 datasheet

## Specifications of ISL6530EVAL2

Main PurposeSpecial Purpose DC/DC, DDR Memory SupplyOutputs And Type2, Non-Isolated
Power - Output31.25WVoltage - Output2.5V, 1.25V
Current - Output10A, 5AVoltage - Input4.5 ~ 5.5V
Regulator TopologyBuckFrequency - Switching300kHz
Board TypeFully PopulatedUtilized Ic / PartISL6530
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Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
V
- V
V
IN
OUT
OUT
∆V
∆I =
x
OUT
f
x L
V
s
IN
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6530 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
equations give the approximate response time interval for
application and removal of a transient load:
L x I
TRAN
t
=
t
=
RISE
FALL
V
- V
IN
OUT
where: I
is the transient load current step, t
TRAN
response time to the application of load, and t
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q
small ceramic capacitors physically close to the MOSFETs
and between the drain of Q
and the source of Q
1
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
13
ISL6530
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
I
RMS
MAX
= ∆I x ESR
For a through-hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge currentrating. These capacitors
must be capable of handling the surge-current at power-up.
Some capacitor series available from reputable manufacturers
are surge current tested.
MOSFET Selection/Considerations
The ISL6530 requires two N-Channel power MOSFETs for
each PWM regulator. These should be selected based upon
r
DS(ON)
requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
L x I
TRAN
switching losses seen when sinking current. The V
V
OUT
regulator will only source current while the V
sink and source. When sourcing current, the upper MOSFET
is the
RISE
realizes most of the switching losses. The lower switch realizes
is the
FALL
most of the switching losses when the converter is sinking
current (see the equations below). These equations assume
linear voltage-current transitions and do not adequately model
power loss due the reverse-recovery of the upper and lower
MOSFET’s body diode. The gate-charge losses are dissipated
by the ISL6530 and don't heat the MOSFETs. However, large
gate-charge increases the switching interval, t
increases the MOSFET switching losses.
LOSSES WHILE SOURCING CURRENT
P
turns on. Place the
UPPER
1
P
LOWER
.
2
LOSSES WHILE SINKING CURRENT
P
UPPER
P
LOWER
Where: D is the duty cycle = V
V
V
V
1
2
OUT
IN
OUT
×
×
I
------------- -
----- -
---------------------------- -
=
+
OUT
×
V
12
L
f
MAX
IN
s
, gate supply requirements, and thermal management
TT
SW
1
2
×
×
×
×
Io
r
D
-- - Io
V
t
=
+
(
)
DS ON
IN
SW
2
2
= Io
x r
x (1 - D)
DS(ON)
2
= Io
x r
x D
DS(ON)
1
2
×
×
(
)
×
Io
r
1 D
-- - Io
V
=
+
(
)
DS ON
2
/ V
,
OUT
IN
t
is the combined switch ON and OFF time, and
SW
f
is the switching frequency.
s
V
2
OUT
×
------------- -
V
IN
DDQ
regulator can
which
×
f
s
×
×
t
f
IN
SW
s
FN9052.2
November 15, 2004