MC33411B Motorola, MC33411B Datasheet - Page 32

• Manufacturer Part Number: MC33411B
• Description: (MC33411A/B) ANALOG CORDLESS PHONE BASEBAND
• Manufacturer: Motorola

calculated. The choice of Q p determines the stability of the
loop. In general, choosing a phase margin of 45 degrees is a
good choice to start calculations. Choosing lower phase
margins will provide somewhat faster lock–times, but also
generate higher overshoots on the control line to the VCO.
This will present a less stable system. Larger values of phase
margin provide a more stable system, but also increase
lock–times. The practical range for phase margin is 30
degrees up to 70 degrees.
lock–time. Since it is quite complicated to accurately
calculate lock time, a good first order approach is:
time. It does not clearly define what the exact frequency
difference is from the desired frequency and it does not show
the effect of phase margin. It assumes, however, that the
phase detector steps up to the desired control voltage
without hesitation. In practice, such step response approach
is not really valid. If the two input frequencies are not locked,
their phase maybe momentarily zero and force the phase
detector into a high impedance mode. Hence, the lock times
may be found to be somewhat higher.
frequency in order for the filter to provide sufficient
attenuation at that frequency. In some applications, the
reference frequency might represent the spacing between
channels. Any feedthrough to the VCO that shows up as a
spur might affect adjacent channel rejection. In theory, with
the loop in lock, there is no signal coming from the phase
detector. But in practice small current pulses and leakage
currents will be supplied to both the VCO and the phase
detector. The external capacitors may show some leakage,
too. Hence, the lower p , the better the reference frequency
is filtered, but the longer it takes for the loop to lock.
0 dB), and thus the absolute value of the complex open loop
gain as shown in equation (6) solves C1:
32
By choosing a value for
The selection of
Equation (12) only provides an order of magnitude for lock
In general, p should be chosen far below the reference
As shown in Figure 48, the open loop gain at p is 1 (or
With C1 known, and equation (5) solve C2 and R2:
C1
+
K
C2
w
pd
2 K
p is strongly related to the desired
T_lock
+
K o T1
R2
n T2
C1 T2
+
p and Q p , T1 and T2 can be
[
T1
C2
T2
w
*
3
p
1
1
1
w
w
p T2
p T1
2
2
MC33411A/B
(12)
(13)
(14)
(15)
external inductor and the frequency required. The free
running frequency of the VCO is determined by:
represents the total capacitance (including internal
capacitance) in parallel with the inductor. The VCO gain can
be easily calculated via the internal varicap transfer curve
shown in Figure 43.
changes 2.0 pF over the voltage range from 1.0 V to 3.0 V:
by:
adequate performance for most applications, an extra pole
may be added for additional reference frequency filtering.
Given that the channel spacing is based on the reference
frequency, and any feedthrough to the first LO may effect
p a r a m e t e r s l i k e a d j a c e n t c h a n n e l r e j e c t i o n a n d
intermodulation. Figure 49 shows a loopfilter architecture
incorporating an additional pole.
the cut–off frequency must be much lower than the reference
frequency. However, it must also be higher than p in order
not to compromise phase margin too much. The following
equations were derived in a similar manner as for the basic
filter previously described.
The VCO gain is dependent on the selection of the
In which L represents the external inductor value and C T
As can be derived from Figure 43, the varicap capacitance
Combining (16) with (17) the VCO gain can be determined
Although the basic loopfilter previously described provides
For the additional pole formed by R3 and C3 to be efficient,
K
o
+
with Additional Integrating Element
j2.0V
From
Phase
Detector
1
Figure 49. Loop Filter
2
p
1
Cvar
f
LC
MOTOROLA RF/IF DEVICE DATA
+
C1
T
2
*
+
p
2
1
2.0 pF
p
R2
C2
2.0 V
LC
R3
T
L C
T
1
C3
To VCO
Cvar
2
(16)
(17)
(18)