isl6520acrz-t Intersil Corporation, isl6520acrz-t Datasheet - Page 8

• Manufacturer Part Number: isl6520acrz-t
• Description: Single Synchronous Buck Pulse-width Modulation Pwm Controller Isl6520a
• Manufacturer: Intersil Corporation

Compensation Break Frequency Equations
Figure 6 shows an asymptotic plot of the DC/DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak due to the high Q factor of the output filter and is not
shown in Figure 6. Using the above guidelines should give a
Compensation Gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
Check the compensation gain at F
the error amplifier. The Closed Loop Gain is constructed on
the graph of Figure 6 by adding the Modulator Gain (in dB) to
the Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
compensation transfer function and plotting the gain.
The compensation gain uses external impedance networks
Z
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
F
F
5. Place 2
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
FB
Z1
Z2
100
-20
-40
-60
80
60
40
20
0
and Z
=
=
----------------------------------- -
2π x R
------------------------------------------------------ -
2π x R
10
20LOG
(R
2
MODULATOR
IN
/R
ND
(
1
1
to provide a stable, high bandwidth (BW) overall
)
GAIN
2
100
1
Pole at Half the Switching Frequency.
x C
1
+
R
1
3
) x C
1K
F
Z1
F
FREQUENCY (Hz)
3
LC
F
Z2
10K
F
F
8
F
P1
P2
F
P1
ESR
(V
P2
=
=
100K
IN
20LOG
F
-------------------------------------------------------- -
2π x R
----------------------------------- -
2π x R
/DV
with the capabilities of
P2
OSC
OPEN LOOP
ERROR AMP GAIN
1
1M
2
3
)
x
x C
1
COMPENSATION
⎛
⎜
⎝
CLOSED LOOP
C
--------------------- -
C
3
1
1
10M
GAIN
x C
+
GAIN
C
2
2
(EQ. 5)
⎞
⎟
⎠
ISL6520A
Modern components and loads are capable of producing
transient load rates above 1A/ns. High frequency capacitors
initially supply the transient and slow the current load rate
seen by the bulk capacitors. The bulk filter capacitor values
are generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6520A will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
ΔI =
V
IN
Fs x L
- V
OUT
x
V
V
OUT
IN
ΔV
OUT
= ΔI x ESR
March 28, 2007
FN9016.5
(EQ. 6)