HIP6014 Intersil Corporation, HIP6014 Datasheet - Page 9

• Manufacturer Part Number: HIP6014
• Description: Buck and Synchronous-Rectifier (PWM) Controller and Output Voltage Monitor
• Manufacturer: Intersil Corporation

suitable component. In most cases, multiple electrolytic
capacitors of small case size perform better than a single
large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
HIP6014 will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
where: I
response time to the application of load, and t
response time to the removal of load. With a +5V input
source, the worst case response time can be either at the
application or removal of load and dependent upon the
DACOUT setting. Be sure to check both of these equations
at the minimum and maximum output levels for the worst
case response time. With a +12V input, and output voltage
level equal to DACOUT, t
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk
capacitors to supply the current needed each time Q
on. Place the small ceramic capacitors physically close to
the MOSFETs and between the drain of Q
of Q
t
RISE
I =
2
.
=
V
IN
TRAN
Fs x L
V
- V
L x I
IN
OUT
- V
is the transient load current step, t
TRAN
OUT
V
V
OUT
IN
FALL
t
FALL
9
is the longest response time.
V
OUT
=
L x I
=
V
OUT
TRAN
I x ESR
1
and the source
FALL
RISE
is the
1
is the
turns
HIP6014
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.
For a through hole design, several electrolytic capacitors
(Panasonic HFQ series or Nichicon PL series or Sanyo MV-
GX or equivalent) may be needed. For surface mount
designs, solid tantalum capacitors can be used, but caution
must be exercised with regard to the capacitor surge current
rating. These capacitors must be capable of handling the
surge-current at power-up. The TPS series available from
AVX, and the 593D series from Sprague are both surge
current tested.
MOSFET Selection/Considerations
The HIP6014 requires 2 N-Channel power MOSFETs.
These should be selected based upon r
requirements, and thermal management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss
components; conduction loss and switching loss. The
conduction losses are the largest component of power
dissipation for both the upper and the lower MOSFETs.
These losses are distributed between the two MOSFETs
according to duty factor (see the equations below). Only the
upper MOSFET has switching losses, since the Schottky
rectifier clamps the switching node before the synchronous
rectifier turns on. These equations assume linear voltage-
current transitions and do not adequately model power loss
due the reverse-recovery of the lower MOSFET’s body
diode. The gate-charge losses are dissipated by the
HIP6014 and don't heat the MOSFETs. However, large gate-
charge increases the switching interval, t
the upper MOSFET switching losses. Ensure that both
MOSFETs are within their maximum junction temperature at
high ambient temperature by calculating the temperature
rise according to package thermal-resistance specifications.
A separate heatsink may be necessary depending upon
MOSFET power, package type, ambient temperature and air
flow.
P
Where: D is the duty cycle = V
P
LOWER
UPPER
t
F
SW
= Io
S
= Io
is the switching frequency.
is the switch ON time, and
2
2
x r
x r
DS(ON)
DS(ON)
x D +
x (1 - D)
OUT
/ V
1
2
IN
Io x V
,
DS(ON)
IN
SW
x t
which increases
SW
, gate supply
x F
S