ISL6227CAZ Intersil, ISL6227CAZ Datasheet - Page 23

• Manufacturer Part Number: ISL6227CAZ
• Description: IC CONTROLLER DDR, DDR2 28QSOP
• Manufacturer: Intersil
• Type: Pulse Width Modulator Controllerr

Specifications of ISL6227CAZ

• Applications: Controller, DDR, DDR2
• Voltage - Input: 5 ~ 28 V
• Number Of Outputs: 2
• Voltage - Output: 0.9 ~ 5.5 V
• Operating Temperature: -10°C ~ 100°C
• Mounting Type: Surface Mount
• Package / Case: 28-QSOP
• Current, Output: 10 mA
• Current, Supply: 1.8 mA
• Frequency, Oscillator: 300 kHz
• Package Type: QSOP-28
• Regulator Type: DC-DC
• Voltage, Input: 5 V
• Voltage, Output: 5.5 V
• Voltage, Supply: 5 V
• Peak Reflow Compatible (260 C): Yes
• Rohs Compliant: Yes
• Lead Free Status / RoHS Status: Lead free / RoHS Compliant

MOSFET and the source terminal of the lower MOSFET, in
order to clamp the parasitic voltage ringing at the phase
node in switching.
Choosing MOSFETs
For a notebook battery with a maximum voltage of 28V, at
least a minimum 30V MOSFETs should be used. The design
has to trade off the gate charge with the r
MOSFET:
• For the lower MOSFET, before it is turned on, the body
• In the turning off process of the lower MOSFET, the load
This results in much less switching loss of the lower
MOSFETs.
The duty cycle is often very small in high battery voltage
applications, and the lower MOSFET will conduct most of
the switching cycle; therefore, the lower the r
lower MOSFET, the less the power loss. The gate charge for
this MOSFET is usually of secondary consideration.
The upper MOSFET does not have this zero voltage
switching condition, and because it conducts for less time
compared to the lower MOSFET, the switching loss tends to
be dominant. Priority should be given to the MOSFETs with
less gate charge, so that both the gate driver loss, and
switching loss, will be minimized.
For the lower MOSFET, its power loss can be assumed to be
the conduction loss only.
For the upper MOSFET, its conduction loss can be written as
Equation 27:
and its switching loss can be written as Equation 28:
The peak and valley current of the inductor can be obtained
based on the inductor peak-to-peak current and the load
current. The turn-on and turn-off time can be estimated with the
given gate driver parameters in the “Electrical Specifications”
Table on page 3. For example, if the gate driver turn-on path of
MOSFET has a typical on-resistance of 4W, its maximum
turn-on current is 1.2A with 5V VCC. This current would decay
as the gate voltage increased. With the assumption of linear
current decay, the turn-on time of the MOSFETs can be written
with Equation 29:
P
P
P
t
on
lower
uppercond
uppersw
diode has been conducting. The lower MOSFET driver will
not charge the miller capacitor of this MOSFET.
current will shift to the body diode first. The high dv/dt of
the phase node voltage will charge the miller capacitor
through the lower MOSFET driver sinking current path.
=
---------------- -
I
(
2Q
driver
V
IN
(
gd
V
)
(
IN
≈
V
IN
(
)
1 D V
=
)
–
=
V
------------------------------------------- -
IN
(
D V
(
I
IN
vally
IN
)
)I
2
)I
load
T
load
on
23
f
2
sw
r
2
DS ON
r
DS ON
+
(
V
-------------------------------------------- -
(
IN
)Lower
I
)upper
peak
DS(ON)
2
T
off
DS(ON)
f
sw
of the
of the
(EQ. 26)
(EQ. 29)
(EQ. 27)
(EQ. 28)
ISL6227
Q
voltage has fallen to zero, it gets charged. Similarly, the turn-off
time can be estimated based on the gate charge and the gate
drivers sinking current capability.
The total power loss of the upper MOSFET is the sum of the
switching loss and the conduction loss. The temperature rise on
the MOSFET can be calculated based on the thermal
impedance given on the datasheet of the MOSFET. If the
temperature rise is too much, a different MOSFET package
size, layout copper size, and other options have to be
considered to keep the MOSFET cool. The temperature rise
can be calculated by Equation 30:
The MOSFET gate driver loss can be calculated with the
total gate charge and the driver voltage V
MOSFET only charges the miller capacitor at turn-off.
Based on Equation 31, the system efficiency can be
estimated by the designer.
Confining the Negative Phase Node Voltage Swing
with Schottky Diode
At each switching cycle, the body diode of the lower MOSFET
will conduct before the MOSFET is turned on, as the inductor
current is flowing to the output capacitor. This will result in a
negative voltage on the phase node. The higher the load
current, the lower this negative voltage. This voltage will ring
back less negative when the lower MOSFET is turned on.
A total 400ns period is given to the current sample-and-hold
circuit on the ISEN pin to sense the current going through the
lower MOSFET after the upper MOSFET turns off. An
excessive negative voltage on the lower MOSFET will be
treated as overcurrent. In order to confine this voltage, a
schottky diode can be used in parallel with the lower MOSFET
for high load current applications. PCB layout parasitics should
be minimized in order to reduce the negative ringing of phase
voltage.
The second concern for the phase node voltage going into
negative is that the boot strap capacitor between the BOOT
and PHASE pin could get be charged higher than VCC voltage,
exceeding the 6.5V absolute maximum voltage between BOOT
and PHASE when the phase node voltage became negative. A
resistor can be placed between the cathode of the boot strap
diode and BOOT pin to increase the charging time constant of
the boot cap. This resistor will not affect the turn-on and off of
the upper MOSFET.
Schottky diode can reduce the reverse recovery of the lower
MOSFET when transition from freewheeling to blocking,
therefore, it is generally good practice to have a Schottky
diode closely parallel with the lower MOSFET. B340LA, from
Diodes, Inc.®, can be used as the external Schottky diode.
T
P
rise
gd
driver
is used because when the MOSFET drain-to-source
=
=
θ P
ja totalpower loss
V
cc
Q
gs
F
sw
CC
. The lower
May 4, 2009
(EQ. 30)
(EQ. 31)
FN9094.7