CA3318CE Intersil, CA3318CE Datasheet - Page 8

• Manufacturer Part Number: CA3318CE
• Description: 8 BIT "FLASH" A/D
• Manufacturer: Intersil

Specifications of CA3318CE

• Rohs Status: RoHS non-compliant
• Other names: CA3318

At the same time a second set of commutating capacitors
and amplifiers is also auto-balanced. The balancing of the
second-stage amplifier at its intrinsic trip point removes any
tracking differences between the first and second amplifier
stages. The cascaded auto-balance (CAB) technique, used
here, increases comparator sensitivity and temperature
tracking.
In the “Sample Unknown” phase, all ladder tap switches and
comparator shorting switches are opened. At the same time
V
Since the other end of the capacitors are now looking into an
effectively open circuit, any input voltage that differs from the
previous tap voltage will appear as a voltage shift at the
comparator amplifiers. All comparators that had tap voltages
greater than V
comparators that had tap voltages lower than V
“low” state.
The status of all these comparator amplifiers is AC coupled
through the second-stage comparator and stored at the end
of this phase (φ2) by a latching amplifier stage. The latch
feeds a second latching stage, triggered at the end of φ1.
This delay allows comparators extra settling time. The status
of the comparators is decoded by a 256 to 9-bit decoder
array, and the results are clocked into a storage register at
the end of the next φ2.
A 3-stage buffer is used at the output of the 9 storage regis-
ters which are controlled by two chip-enable signals. CE1
will independently disable B1 through B6 when it is in a high
state. CE2 will independently disable B1 through B8 and the
OF buffers when it is in the low state.
To facilitate usage of this device, a phase control input is
provided which can effectively complement the clock as it
enters the chip.
Continuous-Clock Operation
One complete conversion cycle can be traced through the
CA3318 via the following steps. (Refer to timing diagram.)
With the phase control in a “low” state, the rising edge of the
clock input will start a “sample” phase. During this entire
“high” state of the clock, the comparators will track the input
voltage and the first-stage latches will track the comparator
outputs. At the falling edge of the clock, all 256 comparator
outputs are captured by the 256 latches. This ends the “sam-
ple” phase and starts the “auto-balance” phase for the com-
parators. During this “low” state of the clock, the output of
the latches settles and is captured by a second row of
latches when the clock returns high. The second-stage latch
output propagates through the decode array, and a 9-bit
code appears at the D inputs of the output registers. On the
next falling edge of the clock, this 9-bit code is shifted into
the output registers and appears with time delay t
data at the output of the three-state drivers. This also marks
the end of the next “sample” phase, thereby repeating the
conversion process for this next cycle.
lN
is switched to the first set of commutating capacitors.
lN
will go to a “high” state at their outputs. All
lN
will go to a
D
as valid
CA3318
8
Pulse-Mode Operation
The CA3318 needs two of the same polarity clock edges to
complete a conversion cycle: If, for instance, a negative
going clock edge ends sample “N”, then data “N” will appear
after the next negative going edge. Because of this require-
ment, and because there is a maximum sample time of
500ns (due to capacitor droop), most pulse or intermittent
sample applications will require double clock pulsing.
If an indefinite standby state is desired, standby should be in
auto-balance, and the operation would be as in Figure 3A.
If the standby state is known to last less than 500ns and
lowest average power is desired, then operation could be as
in Figure 3B.
Increased Accuracy
In most cases the accuracy of the CA3318 should be
sufficient without any adjustments. In applications where
accuracy is of utmost importance, five adjustments can be
made to obtain better accuracy, i.e., offset trim; gain trim;
and
Offset Trim
In general, offset correction can be done in the preamp
circuitry by introducing a DC shift to V
of the op amp. When this is not possible the V
be adjusted to produce an offset trim. The theoretical input
voltage to produce the first transition is
tion is as follows:
If V
single-turn 50Ω pot connected between V
will accomplish the adjustment. Set V
the pot until the 0-to-1 transition occurs.
If V
then the 50Ω pot should be connected between V
negative voltage of about 2 LSBs. The trim procedure is as
stated previously.
Gain Trim
In general, the gain trim can also be done in the preamp
circuitry by introducing a gain adjustment for the op amp.
When this is not possible, then a gain adjustment circuit
should be made to adjust the reference voltage. To perform
this trim, V
That voltage is
follows:
To perform the gain trim, first do the offset trim and then
apply the required V
adjust V
V
V
lN
lN
lN
lN
1
(0 to 1 transition) =
(255 to 256 transition) = V
/
for the first transition is less than the theoretical, then a
4
for the first transition is greater than the theoretical,
,
REF
1
/
2
lN
and
+ until that transition occurs on the outputs.
should be set to the 255 to overflow transition.
1
/
3
3
/
4
LSB less than V
lN
point trim.
= V
for the 255 to overflow transition. Now
1
/
REF
2
= V
LSB =
/512.
REF
REF
REF
1
(511/512).
/
- V
2
lN
lN
(V
+ and is calculated as
REF
1
to 1/2 LSB and trim
or by the offset trim
REF
/
2
REF
/512
LSB. The equa-
/256)
REF
- and ground
REF
- input can
- and a