OPA687 Burr-Brown, OPA687 Datasheet - Page 12

• Manufacturer Part Number: OPA687
• Description: Wideband / Ultra-Low Noise / Voltage Feedback OPERATIONAL AMPLIFIER With Power Down
• Manufacturer: Burr-Brown

FIGURE 6. Op Amp Noise Analysis Model.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO MINIMIZE NOISE
The OPA687 provides a very low input noise voltage while
requiring a low 18.5mA of quiescent current. To take full
advantage of this low input noise, careful attention to the
other possible noise contributors is required. Figure 6 shows
the op amp noise analysis model with all the noise terms
included. In this model, all the noise terms are taken to be
noise voltage or current density terms in either nV/ Hz or
pA/ Hz.
The total output spot-noise voltage can be computed as the
square root of the squared contributing terms to the output
noise voltage. This computation is adding all the contribut-
ing noise powers at the output by superposition, then taking
the square root to get back to a spot-noise voltage. Equation
1 shows the general form for this output noise voltage using
the terms shown in Figure 6.
Equation 1
Dividing this expression by the noise gain (NG = 1 + R
will give the equivalent input-referred, spot-noise voltage at
the non-inverting input as shown in Equation 2.
Equation 2
Putting high resistor values into Equation 2 can quickly
dominate the total equivalent input-referred noise. A source
impedance on the non-inverting input of 56
Johnson voltage noise term equal to just that for the ampli-
fier itself. Holding the gain and source resistors low (as was
used in the Typical Performance Curves) will minimize the
resistor noise contribution in Equation 2. Evaluating Equa-
E
E
E
O
RS
N
R
S
E
E
4kT
4kTR
R
NI
G
NI
2
®
S
I
I
OPA687
BN
BN
I
BN
R
E
NI
S
R
2
S
4
R
kTR NG
G
kTR
OPA687
S
S
I
BI
2
R
F
4kT = 1.6E –20J
I
I R
BI
BI
NG
at 290 K
R
4kTR
F
F
2
F
will add a
4
kTR NG
kTR
NG
F
F
/R
E
O
F
G
)
12
tion 2 for the circuit of Figure 1 will give a total equivalent
input noise of 1.4nV/ Hz. This is slightly increased from the
0.95nV/ Hz for the op amp itself due to the contribution of
the resistor and bias current noise terms.
FREQUENCY RESPONSE CONTROL
Voltage feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90 , as
it does in high gain configurations. At low gains (increased
feedback factors), most high-speed amplifiers will exhibit a
more complex response with lower phase margin. The
OPA687 is compensated to give a maximally flat 2nd-order
Butterworth closed-loop response at a non-inverting gain of
+20 (Figure 1). This results in a typical gain of +20 band-
width of 290MHz, far exceeding that predicted by dividing
the 3600MHz GBP by 20. Increasing the gain will cause the
phase margin to approach 90 and the bandwidth to more
closely approach the predicted value of (GBP/NG). At a
gain of +50, the OPA687 will very nearly match the 72MHz
bandwidth predicted using the simple formula and the typi-
cal GBP of 3600MHz.
Inverting operation offers some interesting opportunities to
increase the available gain bandwidth product. When the
source impedance is matched by the gain resistor (Figure 2),
the signal gain is (1 + R
bandwidth purposes is (1 + R
almost in half, increasing the minimum stable gain for
inverting operation under these condition to –20 and the
equivalent gain bandwidth product to 7.2GHz.
DRIVING CAPACITIVE LOADS
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an A/D converter, including
additional external capacitance which may be recommended
to improve A/D linearity. A high-speed, high open-loop
gain amplifier like the OPA687 can be very susceptible to
decreased stability and closed-loop response peaking when
a capacitive load is placed directly on the output pin. When
the amplifier’s open-loop output resistance is considered,
this capacitive load introduces an additional pole in the
signal path that can decrease the phase margin. Several
external solutions to this problem have been suggested.
When the primary considerations are frequency response
flatness, pulse response fidelity and/or distortion, the sim-
plest and most effective solution is to isolate the capacitive
load from the feedback loop by inserting a series isolation
resistor between the amplifier output and the capacitive
load. This does not eliminate the pole from the loop re-
sponse, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag
from the capacitive load pole, thus increasing the phase
margin and improving stability.
F
/R
F
/2R
G
) while the noise gain for
G
). This cuts the noise gain