OPA687 Burr-Brown, OPA687 Datasheet - Page 9

• Manufacturer Part Number: OPA687
• Description: Wideband / Ultra-Low Noise / Voltage Feedback OPERATIONAL AMPLIFIER With Power Down
• Manufacturer: Burr-Brown

FIGURE 3. Wideband, High Sensitivity, OC-3
required for application as a transimpedance amplifier. Fig-
ure 3 shows one possible transimpedance design example
that would be particularly suitable for the 155Mbit data rate
of an OC-3 receiver. Designs that require high bandwidth
from a large area detector with relatively low transimpedance
gain will benefit from the low input voltage noise for the
OPA687. The amplifier’s input voltage noise is peaked up,
at the output, over frequency by the diode source capaci-
tance and can, in many cases, become the dominant output
noise contribution. The key elements to the design are the
expected diode capacitance (C
age (–V
the GBP for the OPA687 (3600MHz). With these three
variables set (and including the parasitic input capacitance
for the OPA687 added to C
(C
To achieve a maximally flat 2nd-order Butterworth fre-
quency response, the feedback pole should be set to:
Adding the common-mode and differential-mode input ca-
pacitance (1.2 + 2.5)pF to the 1pF diode source capacitance
of Figure 3 (C
using the 3600MHz GBP for the OPA687, will require a
feedback pole set to 71MHz to get a maximum bandwidth
design. This will require a total feedback capacitance of
0.16pF.
Using this maximum bandwidth, maximally flat frequency
response target will give an approximate –3dB bandwidth
set by:
F
) may be set to control the frequency response.
B
100pF
) applied, the desired transimpedance gain, R
–V
1/(2 R
D
B
), and targeting a 12k transimpedance gain
f
Transimpedance Amplifier.
–3dB
Photodiode
0.1 F
1pF
F
C
= (GBP/2 R
F
) = (GBP/(4 R
12k
D
), the feedback capacitor value
D
) with the reverse bias volt-
OPA687
+5V
–5V
F
C
Supply Decoupling
D
)Hz
0.16pF
12k
F
Not Shown
R
C
C
F
F
D
))
F
, and
9
The example of Figure 3 will give approximately 100MHz
flat bandwidth using the 0.16pF feedback compensation
capacitor. This bandwidth will easily support an OC-3 re-
ceiver with exceptional sensitivity.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:
Where:
i
i
e
C
f
Evaluating this expression up to the feedback pole frequency
at 71MHz for the circuit of Figure 3, gives an equivalent
input noise current of 3.0pA/ Hz. This is somewhat higher
than the 2.5pA/ Hz for just the op amp itself. This total
equivalent input current noise is being slightly increased by
the last term in the equivalent input noise expression. It is
essential in this case to use a low voltage noise op amp. For
example, if a slightly higher input noise voltage, but other-
wise identical, op amp were used instead of the OPA687 in
this application (say 2.0nV/ Hz), the total input-referred
current noise would increase to 4.0pA/ Hz. Low input
voltage noise is required for the best sensitivity in these
wideband transimpedance applications. This is often un-
specified for dedicated transimpedance amplifiers with a
total output noise for a specified source capacitance given
instead. It is the relatively high input voltage noise for those
components that cause higher than expected output noise if
the source capacitance is higher than expected.
LOW GAIN COMPENSATION FOR IMPROVED SFDR
A new external compensation technique may be used at low
signal gains to retain the full slew rate and noise benefits of
the OPA687, while maintaining the increased loop gain and
the associated improvement in distortion offered by the
decompensated architecture. This technique shapes the loop
gain for good stability while giving an easily controlled
second-order low pass frequency response. This technique
was used for the circuit on the front page of the data sheet
in a differential configuration to achieve extremely high
SFDR through high frequencies. That circuit is set up for a
differential gain of 8.5V/V from a differential input signal to
the output. Using the transformer shown will improve the
noise figure and translate from a single to a differential
EQ
N
N
D
= Equivalent input noise current if the output noise is
= Input current noise for the op amp inverting input
= Input voltage noise for the op amp
= Total Inverting Node Capacitance
= Bandlimiting frequency in Hz (usually a post filter
bandlimited to f < 1/(2 R
prior to further signal processing)
i
EQ
i
2
N
4
R
kT
F
R
e
N
F
OPA687
F
2
C
D
)
e
N
2
3
C f
D
2
®