MAX17582GTM+ Maxim Integrated Products, MAX17582GTM+ Datasheet - Page 40

• Manufacturer Part Number: MAX17582GTM+
• Description: IC PWM CTRLR STP-DN DL 48TQFN
• Manufacturer: Maxim Integrated Products
• Series: Quick-PWM™r

Specifications of MAX17582GTM+

• Applications: Controller, Intel IMVP-6.5™
• Voltage - Input: 4.5 ~ 5.5 V
• Number Of Outputs: 1
• Voltage - Output: 0.01 ~ 1.5 V
• Operating Temperature: -40°C ~ 105°C
• Mounting Type: Surface Mount
• Package / Case: 48-TQFN Exposed Pad
• Lead Free Status / RoHS Status: Lead free / RoHS Compliant

Dual-Phase, Quick-PWM Controller for
IMVP-6.5 CPU Core Power Supplies
side. According to the manufacturer’s data sheet, a sin-
gle IRF7811W has a maximum gate charge of 24nC
(V
boost capacitance would be:
Selecting the closest standard value, this example
requires a 0.22μF ceramic capacitor.
The current-balance compensation capacitor (C
integrates the difference between the main and sec-
ondary current-sense voltages. The internal compensa-
tion resistor (R
response by increasing the phase margin. This allows
the dynamics of the current-balance loop to be opti-
mized. Excessively large capacitor values increase the
integration time constant, resulting in larger current dif-
ferences between the phases during transients.
Excessively small capacitor values allow the current
loop to respond cycle-by-cycle, but can result in small
DC current variations between the phases. For most
applications, a 470pF capacitor from CCI to the switch-
ing regulator’s output works well.
Connecting the compensation network to the output
(V
voltage signal, especially during transients.
Voltage positioning dynamically lowers the output volt-
age in response to the load current, reducing the out-
put capacitance and processor’s power-dissipation
requirements. The controller uses a transconductance
amplifier to set the transient and DC output-voltage
droop (Figure 2) as a function of the load. This adjusta-
bility allows flexibility in the selected current-sense
resistor value or inductor DCR, and allows smaller cur-
rent-sense resistance to be used, reducing the overall
power dissipated.
Connect a resistor (R
the DC steady-state droop (load line) based on the
required voltage-positioning slope (R
where the effective current-sense resistance (R
depends on the current-sense method (see the Current
Sense section), and the voltage-positioning amplifier’s
40
OUT
GS
______________________________________________________________________________________
= 5V). Using the above equation, the required
) allows the controller to feed-forward the output-
Current-Balance Compensation (CCI)
C
BST_
CCI
R
FB
Steady-State Voltage Positioning
=
FB
=
= 200kΩ) improves transient
2 24
R
) between FB and V
200
Voltage Positioning and
×
SENSE m FB
R
mV
DROOP
nC
Loop Compensation
G
=
(
0 24
.
DROOP
)
μF
):
OUT
SENSE
to set
CCI
)
)
transconductance (G
defined in the Electrical Characteristics table. The con-
troller sums together the input signals of the current-
sense inputs (CSP_, CSN_).
When the inductors’ DCR is used as the current-sense
element (R
should include an NTC thermistor to minimize the tem-
perature dependence of the voltage-positioning slope.
The output-voltage-adjustable range for continuous-
conduction operation is restricted by the nonadjustable
minimum off-time one-shot and the number of phases.
For best dropout performance, use the slower (200kHz)
on-time settings. When working with low input voltages,
the duty-factor limit must be calculated using worst-
case values for on- and off-times. Manufacturing toler-
ances and internal propagation delays introduce an
error to the on-times. This error is greater at higher fre-
quencies. Also, keep in mind that transient-response
performance of buck regulators operated too close to
dropout is poor, and bulk output capacitance must
often be added (see the V
Multiphase Quick-PWM Design Procedure section).
The absolute point of dropout is when the inductor cur-
rent ramps down during the minimum off-time (ΔI
as much as it ramps up during the on-time (ΔI
ratio h = ΔI
slew the inductor current higher in response to
increased load, and must always be greater than 1. As
h approaches 1, the absolute minimum dropout point,
the inductor current cannot increase as much during
each switching cycle and V
unless additional output capacitance is used.
A reasonable minimum value for h is 1.5, but adjusting
this up or down allows trade-offs between V
capacitance, and minimum operating voltage. For a
given value of h, the minimum operating voltage can be
calculated as:
where η
switching regulators, V
droop, V
drops in the discharge and charge paths (see the On-
Time One-Shot section), and t
Electrical Characteristics table. The absolute minimum
input voltage is calculated with h = 1.
V
IN MIN
Minimum Input-Voltage Requirements
(
DROP1
TOTAL
)
SENSE
UP
=
η
/ΔI
V
TOTAL
DROP
is the total number of out-of-phase
and V
DOWN
= R
2
⎡
⎢
⎢
⎣
and Dropout Performance
DCR
DROP2
-
1
V
m(FB)
is an indicator of the ability to
-
V
FB
DROP
FB
η
), each current-sense input
TOTAL
is the voltage-positioning
-
V
) is typically 600μS as
1
are the parasitic voltage
DROOP
SAG
+
SAG
h t
V
OFF(MIN)
DROOP
×
greatly increases
O O FF MIN SW
equation in the
+
V
(
DROP
is from the
SAG
)
f
1
UP
, output
DOWN
⎤
⎥
⎥
⎦
). The
+
)