LM4781TA/NOPB National Semiconductor, LM4781TA/NOPB Datasheet - Page 18

• Manufacturer Part Number: LM4781TA/NOPB
• Description: IC AMP AUDIO PWR 35W AB TO220-27
• Manufacturer: National Semiconductor
• Series: Overture™r
• Type: Class ABr

Specifications of LM4781TA/NOPB

• Output Type: 3-Channel
• Max Output Power X Channels @ Load: 35W x 3 @ 8 Ohm
• Voltage - Supply: 20 V ~ 70 V, ±10 V ~ 35 V
• Features: Depop, Mute, Short-Circuit and Thermal Protection
• Mounting Type: Through Hole
• Package / Case: TO-220-27 (Bent and Staggered Leads)
• For Use With: LM4781TABD - BOARD EVALUATION LM4781TA
• Lead Free Status / RoHS Status: Lead free / RoHS Compliant
• Other names: *LM4781TA; *LM4781TA/NOPB; LM4781TA

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Application Information
the device employs Under-Voltage Protection, which elimi-
nates unwanted power-up and power-down transients. The
basis for these functions are a stable and constant half-
supply potential. In a split-supply application, ground is the
stable half-supply potential. But in a single-supply applica-
tion, the half-supply needs to charge up at the same rate as
the supply rail, V
clickless and popless turn-on more challenging. Any uneven
charging of the amplifier inputs will result in output clicks and
pops due to the differential input topology of the LM4781.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4781. In Figure 4, the resistor R
serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal perfor-
mance.
As shown in Figure 4, the resistors labeled R
the LM4781 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R
resistors bring up the inputs at the same rate resulting in a
popless turn-on. Adjusting these resistors values slightly
may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components is required to meet
the design targets of an application. The choice of external
component values that will affect gain and low frequency
response are discussed below.
The gain of each amplifier is set by resistors R
non-inverting configuration shown in Figure 1. The gain is
found by Equation (6) below:
For best noise performance, lower values of resistors are
used. A value of 1kΩ is commonly used for R
setting the value of R
the gain should be set no lower than 10V/V and no higher
than 50V/V. Gain settings below 10V/V may experience
instability and using the LM4781 for gains higher than 50V/V
will see an increase in noise and THD.
The combination of R
pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
Equation (7) shown below:
If an input coupling capacitor is used to block DC from the
inputs as shown in Figure 5, there will be another high pass
filter created with the combination of C
using a input coupling capacitor R
bias point on the amplifier’s input terminal. The resulting
-3dB frequency response due to the combination of C
R
With large values of R
outputs when the inputs are left floating. Decreasing the
value of R
IN
can be found from Equation (8) shown below:
IN
or not letting the inputs float will remove the
f
CC
IN
A
f
V
i
= 1 / (2πR
. This makes the task of attaining a
f
= 1 / (2πR
BI
i
IN
= 1 + R
for the desired gain. For the LM4781
with C
, namely 10kΩ and 200kΩ. These
oscillations may be observed on the
i
f
(see Figure 1) creates a high
IN
/ R
i
C
C
i
IN
) (Hz)
i
IN
(V/V)
is needed to set the DC
) (Hz)
IN
(Continued)
and R
BI
f
and R
help bias up
i
IN
and then
. When
i
for the
IN
and
INP
(6)
(7)
(8)
18
oscillations. If the value of R
C
frequency response.
HIGH PERFORMANCE CONSIDERATIONS
Using low cost electrolytic capacitors in the signal path such
as C
performance. However, electrolytic capacitors are less linear
than other premium capacitors. Higher THD+N performance
may be obtained by using high quality polypropylene capaci-
tors in the signal path. A more cost effective solution may be
the use of smaller value premium capacitors in parallel with
the larger electrolytic capacitors. This will maintain signal
quality in the upper audio band where any degradation is
most noticeable while also coupling in the signals in the
lower audio band for good bass response.
Distortion is introduced as the audio signal approaches the
lower -3dB point, determined as discussed in the section
above. By using larger values of capacitors such that the
-3dB point is well outside of the audio band will reduce this
distortion and improve THD+N performance.
Increasing the value of the large supply bypass capacitors
will improve burst power output. The larger the supply by-
pass capacitors the higher the output pulse current without
supply droop increasing the peak output power. This will also
increase the headroom of the amplifier and reduce THD.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated
circuit designs to discrete amplifier designs. Discrete transis-
tor amps often “run out of gain” at high frequencies and
therefore have small bandwidths to noise as indicated below.
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic
distortion and improve frequency response. It is this addi-
tional bandwidth that can lead to erroneous signal-to-noise
measurements if not considered during the measurement
process. In the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10in
bandwidth, (200kHz to 2MHz) can result in a 10dB theoreti-
cal difference in the signal-to-noise ratio (white noise is
proportional to the square root of the bandwidth in a system).
In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
“weighting” filter (Note 19). A “weighting” filter alters the
frequency response in order to compensate for the average
human ear’s sensitivity to the frequency spectra. The weight-
ing filters at the same time provide the bandwidth limiting as
discussed in the previous paragraph.
IN
will need to increase in order to maintain the same -3dB
IN
and C
i
(see Figures 1 - 5) will result in very good
IN
is decreased then the value of
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