ad1890jpz Analog Devices, Inc., ad1890jpz Datasheet - Page 12

• Manufacturer Part Number: ad1890jpz
• Description: Sampleport Stereo Asynchronous Sample Rate Converters
• Manufacturer: Analog Devices, Inc.

AD1890/AD1891
Cutoff Frequency Modification
The final important operating concept of the ASRCs is the mod-
ification of the filter cutoff frequency when the output sample
rate (F
during downsampling operation. The AD1890/AD1891 auto-
matically reduces the polyphase filter cutoff frequency under
this condition. This lowering of the cutoff frequency (i.e., the
reduction of the input signal bandwidth) is required to avoid
alias distortion. The AD1890/AD1891 SoundPorts take advan-
tage of the scaling property of the Fourier transform which can
be stated as follows: if the Fourier transform of f(t) is F(w), then
the Fourier transform of f(k
used to linearly compress the frequency response of the filter,
simply by multiplying the coefficient ROM addresses (shown in
Figure 6) by the ratio of F
than F
tion because the time scale of the interpolated signal is so dense
(300 ps resolution) with respect to the cutoff frequency that the
discrete-time representation is a close approximation to the con-
tinuous time function.
The cutoff frequency (–3 dB down) of the FIR filter during
downsampling is given by the following relation:
Downsampling Cutoff Frequency = (F
The AD1890/AD1891 frequency response compression circuit
includes a first order low-pass filter to smooth the filter cutoff
frequency selection during dynamic sample rate conditions.
This allows the ASRC to avoid objectionable clicking sounds
that would otherwise be imposed on the output while the loop
settles to a new sample rate ratio. Hysteresis is also applied to
the filter selection with approximately 300 Hz of cutoff fre-
quency “noise margin,” which limits the available selection of
cutoff frequencies to those falling on an approximately 300 Hz
frequency grid. Thus if a particular sample frequency ratio was
reached by sliding the output sample frequency up, it is possible
that a filter will be chosen with a cutoff frequency that could dif-
fer by as much as 300 Hz from the filter chosen when the same
sample frequency ratio was reached by sliding the output sample
frequency down. This is necessary to ensure that the filter selec-
tion is stable even with severely jittered input sample clocks.
Note that when the filter cutoff frequency is reduced, the transi-
tion band of the filter becomes narrower since the scaling prop-
erty affects all filter characteristics. The number of FIR filter
taps necessarily increases because there are now a smaller num-
ber of longer length polyphase filters. Nominally, when F
greater than F
than F
128 when the ratio of F
filter taps as a function of sample clock ratio is illustrated in Fig-
ure 8. The natural consequence of this increase in filter taps is
an increase in group delay.
SIN
SIN,
SOUT
. This scaling property works without spectral distor-
the number of taps linearly increase to a maximum of
) drops below the input sample rate (F
SIN
, the number of taps is 64. When F
SOUT
SOUT
, to F
t) is F(w/k). This property can be
to F
SIN
SIN
equals 1:2. The number of
SOUT
whenever F
/44.1 kHz)
SIN
SOUT
SOUT
), i.e.,
is less
20 kHz
SOUT
is less
is
–12–
When the AD1890/AD1891 output sample frequency is higher
than the input sample frequency (i.e., upsampling operation),
the cutoff frequency of the FIR polyphase filter can be greater
than 20 kHz. The cutoff frequency of the FIR filter during
upsampling is given by the following relation:
Upsampling Cutoff Frequency = (F
Noise and Distortion Phenomena
There are three noise/distortion phenomena that limit the per-
formance of the AD1890/AD1891 ASRCs. First, there is
broadband, Gaussian noise which results from polyphase filter
selection quantization. Even though the AD1890/AD1891 have
a large number of polyphase filters (the equivalent of 65,536) to
choose from, the selection is not infinite. Second, there is
narrow-band noise which results from the non-ideal synchroni-
zation of the sample clocks to the system clock MCLK, which
leads to a non-ideal computation of the sample clock ratio,
which leads to a non-ideal polyphase filter selection. This noise
source is narrowband because the digital servo control loop
averages the polyphase filter selection, leading to a strong corre-
lation between selections from output to output. In slow mode,
the selection of polyphase filters is completely unaffected by the
clock synchronization. In fast mode, some narrowband noise
modulation may be observed with very long FFT measure-
ments. This situation is analogous to the behavior of a phase
locked loop when presented with a noisy or jittered input.
Third, there are distortion components that are due to the
non-infinite stopband rejection of the low-pass filter response.
Non-infinite stopband rejection means that some amount of
out-of-band spectral energy will alias into the baseband. The
AD1890/AD1891 performance specifications include the effects
of these phenomena.
Note that Figures 15 through 17 are shown with full-scale input
signals. The distortion and noise components will scale with the
input signal amplitude. In other words, if the input signal is at-
tenuated by –20 dB, the distortion and noise components will
also be attenuated by –20 dB. This dependency holds until the
effects of the 20-bit input quantization are reached.
Figure 8. Number of Filter Taps as a Function of
F
SOUT
/F
128
64
SlN
0.5
SAMPLING
DOWN-
1.0
UPSAMPLING
1.5
SIN
/44.1 kHz)
2.0
F
SOUT
/F
20 kHz
SIN
REV. 0